Sharing operational amplifier between two stages of pipelined ADC and/or two channels of signal processing circuitry

ABSTRACT

A mechanism for discharging parasitic capacitance at an input of an operational amplifier, which is shared between two stages of a pipelined analog-to-digital converter and/or two channels of signal processing circuitry, before the amplifier configuration of the stages/channels is switched. The discharging act occurs when a short reset pulse is generated between two clock phases. The short reset pulse is applied to a switch connected to the operational amplifier input. When the reset pulse closes the switch, a discharge path is created and any parasitic capacitance at the operational amplifier input is discharged through the path. The discharging of the parasitic capacitance substantially mitigates the memory effect and the problems associated with the memory effect.

This application is a continuation of application Ser. No. 11/211,566,filed Aug. 26, 2005, now U.S. Pat. No. 7,148,833, which is herebyincorporated by reference in its entirety.

FIELD OF THE INVENTION

The present invention relates to pipelined analog-to-digital convertersthat share operational amplifiers between two stages of the pipeline andsignal processing circuitry that shares an operational amplifier betweentwo signal processing channels.

BACKGROUND OF THE INVENTION

Analog-to-digital converters (ADCs) are useful components in any circuitor system that interfaces analog and digital circuitry and signals. Oneapplication in which analog-to-digital converters are particularlyuseful includes imaging devices such as CMOS and CCD imagers. Imagerstypically convert light photons into analog image signals. These analogimage signals are converted to digital signals, by an analog-to-digitalconverter, and then processed by an image processor or other processingcircuitry.

There is a desire to increase the speed of the analog-to-digitalconversion process in many applications such as e.g., imagers. As such,many applications use pipelined analog-to-digital converters, whichtypically operate faster than non-pipelined analog-to-digitalconverters. FIG. 1 is an illustration of a conventional N-bit pipelinedanalog-to-digital converter 10. The pipelined analog-to-digitalconverter 10 consists of multiple low resolution (e.g., 1.5 bits) stages12 ₁, 12 ₂, . . . , 12 _(n), . . . 12 _(N−1), each of which comprises anarithmetic unit 20 and a two-level decision circuit 40. The pipelinedanalog-to-digital converter 10 further includes digital correction logic14 for outputting an N-bit digital code representing an input analogsignal.

FIG. 1 illustrates the components of the nth stage 12 _(n) in moredetail. It should be appreciated that the other stages 12 ₁, 12 ₂, . . ., 12 _(N−1) contain the same circuitry as the nth stage 12 _(n). Thearithmetic unit 20 comprises a switching block 22, four additionalswitches 24, 26, 28, 30, a sampling capacitor Cs, a feedback capacitorCf, and an operational amplifier 32. The decision circuit 40 includestwo comparators 42, 44 and an encoder 46.

In operation, the arithmetic unit 20 in the first stage 12 ₁ merelyoperates as a sample and hold circuit. In the other stages 12 ₂, . . . ,12 _(n), . . . 12 _(N−1), the arithmetic unit 20 multiplies the incominganalog signal portion V_(RES(n−1)), often referred to as a “residue,” bya factor of two and subtracts from this product one of three voltages+V_(R), 0, −V_(R), based on the closed switch in the switching block 22.The switches of block 22 are opened/closed based on the decision bitsD_(n−1) from a prior stage (e.g., stage 12 _(n−1)). The new residue isfed into the decision circuit 40, where it is compared with twodifferent reference voltages ¼V_(R), −¼V_(R). The encoder generates andoutputs decision bits D_(n) for the stage 12 _(n). The decision bits foreach of the stages 12 ₁, 12 ₂, . . . , 12 _(n), . . . 12 _(N−1) areprocessed by the digital correction logic 14, which removes anyredundancy and outputs the N-bit digital output code.

As can be seen in FIG. 1, the conventional pipelined analog-to-digitalconverter 10 requires one operational amplifier 32 for each stage 12 ₁,12 ₂, . . . , 12 _(n), . . . 12 _(N−1) in the pipeline. The majority ofthe power of the pipelined analog-to-digital converter 10 is consumed byoperational amplifiers 32. Therefore, minimizing the power consumptionof the operational amplifiers 32 is key to the design of low powerpipelined analog-to-digital converters 10.

FIG. 2 illustrates the timing diagram for two stages STAGE 1, STAGE 2 ofthe FIG. 1 pipelined analog-to-digital converter 10. Non-overlappingclock signals PHI1, PHI2 are used to control the switching circuitrycontained within each stage STAGE 1, STAGE 2 to configure how thesampling and feedback capacitors Cs, Cf and the operational amplifier 32are connected.

FIG. 3 illustrates the operational amplifier configuration of the twostages STAGE 1, STAGE 2 when the second clock signal PHI2 is asserted(i.e., has a high level). As can be seen in FIGS. 2 and 3, the firststage STAGE 1 undergoes a sampling operation while the second stageSTAGE 2 undergoes an amplifying operation. That is, the first stage'sarithmetic unit 20 ₁ is configured such the analog input voltage Vin issampled in the sampling capacitor Cs. The second stage's arithmetic unit20 ₂ is configured in a manner such that the operational amplifier 32amplifies the signal stored in the sampling capacitor Cs and outputs theamplified signal as Vout.

FIG. 4 illustrates the operational amplifier configuration of the twostages STAGE 1, STAGE 2 when the first clock signal PHI1 is asserted(i.e., has a high level). As can be seen in FIGS. 2 and 4, the firststage STAGE 1 undergoes the amplifying operation while the second stageSTAGE 2 undergoes the sampling operation. That is, the first stage'sarithmetic unit 20 ₁ is configured such the signal stored in thesampling capacitor Cs is amplified by the operational amplifier 32. Thesecond stage's arithmetic unit 20 ₂ is configured to sample the outputfrom the first stage STAGE 1 and store it in the stage 2 STAGE 2sampling capacitor Cs.

It can be seen from FIGS. 3 and 4 that during the sampling operations,the operational amplifiers 32 performs no useful function; they justconsume power. This occurs because the operational amplifiers 32 areplaced into an open-loop configuration with their inputs and outputsconnected to known voltage levels. To avoid wasting power during everysampling operation, some analog-to-digital converters share oneoperational amplifier 32 between two adjacent stages STAGE 1, STAGE 2 asus shown in FIGS. 5 and 6.

FIGS. 5 and 6 illustrate a circuit 120 of a pipelined analog-to-digitalconverter in which arithmetic units 20 ₁, 20 ₂ of two pipeline stagesSTAGE 1, STAGE 2 share one operational amplifier 32. The amplifier 32can be shared because the circuit 120 contains six switches S1, S2, S3,S4, S5, S6 that are controlled to connect the sampling and feedbackcapacitors Cs, Cf to the operational amplifier 32 inputs and outputsdifferently depending on the operation being performed.

FIG. 5 illustrates the circuit 120 when the second clock signal PHI2 ofFIG. 2 is asserted. While the second clock signal PHI2 is asserted,switch S1 is closed to connect the analog input voltage Vin to the stage1 arithmetic unit 20 ₁ sampling capacitor Cs. Switches S5 and S6 areclosed in the second stage's arithmetic unit 20 ₂ such that theoperational amplifier 32 amplifies, and outputs as Vout, a signal storedin the stage 2 arithmetic unit 20 ₂ sampling capacitor Cs. The otherswitches S2, S3 and S4 are left open. Thus, as can be seen in FIGS. 2and 5, the first stage STAGE 1 undergoes a sampling operation while thesecond stage STAGE 2 undergoes an amplifying operation, but only oneoperational amplifier 32 is connected and used.

FIG. 6 illustrates the circuit 120 when the first clock signal PHI1 ofFIG. 2 is asserted. While the first clock signal PHI1 is asserted,switches S1, S5 and S6 are open, and switches S2, S3 and S4 are closed.As such, the first stage's arithmetic unit 20 ₁ is configured such thata signal stored in the first stage arithmetic unit 20 ₁ samplingcapacitor Cs is amplified by the operational amplifier 32 and output asVout. The second stage's arithmetic unit 20 ₂ is configured to sampleand store an analog input Vin in the stage 2 STAGE 2 sampling capacitorCs. As can be seen in FIGS. 2 and 6, the first stage STAGE 1 undergoesthe amplifying operation while the second stage STAGE 2 undergoes thesampling operation. Again, only one operational amplifier 32 isconnected and used during these operations.

By sharing an operational amplifier 32 between adjacent two stages STAGE1, STAGE 2, the power consumption of the pipelined analog-to-digitalconverter 10 (FIG. 1) can be reduced by half. However due to the finiteDC gain Ao and input parasitic capacitance Cp of the operationalamplifier 32 (FIG. 7 b), the previous output Vo(k-1) adversely effectsthe present output Vo(k), which is known in the art as the “memoryeffect.” The memory effect can cause a non-linearity in the operationalamplifier 32 and thus, analog-to-digital converter output.

Briefly, the memory effect can be described using the followingequations in reference to FIG. 7 a. Ideally, during sampling, chargeshould be represented as Q=(Cf+Cs)×Vin. From charge conversion, at theamplifying phase, Q=Cf×(Vo−Vx)−Cp×Vx−Cs×Vx=(Cf+Cs)×Vin, where Vx is theinput node voltage of the operational amplifier 32. Because theamplifier has a finite gain Ao, Vo=−Ao×Vx −>Vx=−Vo/Ao. This means thatCf×(Vo+Vo/Ao)+Cp×Vo/Ao+Cs×Vo/Ao=(Cf+Cs)×Vin. Therefore,Vo=Vin×(Cf+Cs)/(Cf+(Cf+Cs+Cp)/Ao), which equals Vin×Gc.

In reality, however, there is charge associated with parasiticcapacitance Cp (due to the memory effect). As such, at the samplingstage, as shown in FIG. 7 b, Q=(Cf+Cs)×Vin(k)−Cp×Vin_err(k), whereVin_err(k) is the memory error associated with the parasitic capacitanceCp. Using just the error term, from charge conversion,Vo(k)=−Vin_err(k)×Cp/(Cf+(Cf+Cs+Cp)/Ao)˜=−Vin_err(k)×Cp/Cf, if Ao islarge enough. For the first and second termsVo(k)=Vin(k)×Gc−Vin_err(k)×Cp/(Cf+(Cf+Cs+Cp)/Ao). Since Vin_err(k) comesfrom the previous output, Vin_err(k)=−Vo(k−1)/Ao=−Gc×Vin(k−1)/Ao.Accordingly,Vo(k)=Vin(k)×Gc+Vin(k−1)×Gc/Ao×Cp/(Cf+(Cf+Cs+Cp)/Ao)=Vin(k)×Gc+Vin(k−1)×Gc×e,where e=1/Ao×Cp/(Cf+(Cf+Cs+Cp)/Ao)˜1/Ao×Cp/Cf. It should be noted thatthe second order error are neglected in the above calculations.

In addition, charge injection and kickback noise from the circuitry addto the memory effect error described above. Reducing the memory effectis a key element in designing a pipelined analog-to-digital converterthat shares operational amplifiers between two pipeline stages.Accordingly, there is a need and desire for a pipelinedanalog-to-digital converter that shares an operational amplifier betweentwo pipeline stages, yet does not suffer from the memory effect and theproblems associated with the memory effect.

It is known to divide signal processing circuitry into multiplechannels. For example, imagers often include multiple readout channelswhere one channel processes a specific set of pixel signals and at leastone other channel processes the remaining sets of pixel signals. FIG. 7c illustrates a two channel processing circuit 150 designed to sampleand hold analog input signals and convert the signals into digitalsignals. As shown in FIG. 7 c, the first channel CHANNEL 1 comprises asample and hold circuit 152 _(a) and multiple analog-to-digital pipelinestages 154 _(a), 156 _(a). Similarly, the second channel CHANNEL 2comprises a sample and hold circuit 152 _(b) and multipleanalog-to-digital pipeline stages 154 _(b), 156 _(b). The sample andhold circuits 152 _(a), 152 _(b) share an operational amplifier 32. Theanalog-to-digital pipeline stages 154 _(a), 154 _(b) share anoperational amplifier 32 as do the other analog-to-digital pipelinestages 156 _(a), 156 _(b).

The devices of the two channels CHANNEL 1, CHANNEL 2 share theoperational amplifiers in a similar manner and with similar timing(e.g., FIG. 2) as the adjacent pipelined analog-to-digital converterstages share the operational amplifiers (as discussed above). That is,the channels switch in or out the amplifier based on the operation beingperformed in that portion of the channel. Thus, although the circuit 150achieves the benefits of reducing the number of operational amplifiers,the circuit 150 also suffers from the memory effect. Accordingly, thereis a need and desire for sharing an operational amplifier between twochannels of a signal processing circuit, yet does not suffer from thememory effect and the problems associated with the memory effect.

SUMMARY OF THE INVENTION

The invention provides a pipelined analog-to-digital converter thatshares an operational amplifier between two pipeline stages, yet doesnot suffer from the memory effect and the problems associated with thememory effect.

The invention also provides for the sharing of an operational amplifierbetween two channels of a signal processing circuit, yet does not sufferfrom the memory effect and the problems associated with the memoryeffect.

The above and other features and advantages are achieved in variousexemplary embodiments of the invention by providing a mechanism fordischarging parasitic capacitance at an input of an operationalamplifier, which is shared between two stages of a pipelinedanalog-to-digital converter and/or two channels of signal processingcircuitry, before the amplifier configuration of the stages/channels isswitched. The discharging act occurs when a short reset pulse isgenerated between two clock phases. The short reset pulse is applied toa switch connected to the operational amplifier input. When the resetpulse closes the switch, a discharge path is created and any parasiticcapacitance at the operational amplifier input is discharged through thepath. The discharging of the parasitic capacitance substantiallymitigates the memory effect and the problems associated with the memoryeffect.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing and other advantages and features of the invention willbecome more apparent from the detailed description of exemplaryembodiments provided below with reference to the accompanying drawingsin which:

FIG. 1 is an illustration of a conventional N-bit pipelinedanalog-to-digital converter;

FIG. 2 is a timing diagram for two stages of the FIG. 1 pipelinedanalog-to-digital converter;

FIG. 3 illustrates the operational amplifier configuration of the twostages of the FIG. 1 pipelined analog-to-digital converter in accordancewith one timing of the FIG. 2 timing diagram;

FIG. 4 illustrates the operational amplifier configuration of the twostages of the FIG. 1 pipelined analog-to-digital converter in accordancewith a second timing of the FIG. 2 timing diagram;

FIG. 5 illustrates a first shared operational amplifier configuration ofthe two stages of the FIG. 1 pipelined analog-to-digital converter inaccordance with one timing of the FIG. 2 timing diagram;

FIG. 6 illustrates a second shared operational amplifier configurationof the two stages of the FIG. 1 pipelined analog-to-digital converter inaccordance with a second timing of the FIG. 2 timing diagram;

FIGS. 7 a and 7 b illustrate by comparison the memory effect that arisesin the shared operational amplifier configuration of the two stages ofthe FIG. 1 pipelined analog-to-digital converter;

FIG. 7 c illustrates a two channel signal processing circuit that sharesoperational amplifiers between respective portions of the channels;

FIG. 8 is an exemplary timing diagram for two stages of pipelinedanalog-to-digital converter constructed in accordance with an exemplaryembodiment of the invention;

FIG. 9 illustrates a first shared operational amplifier configuration oftwo stages of the pipelined analog-to-digital converter constructed inaccordance with an exemplary embodiment of the invention in accordancewith one timing of the FIG. 8 timing diagram;

FIG. 10 illustrates a second shared operational amplifier configurationof two stages of the pipelined analog-to-digital converter constructedin accordance with an exemplary embodiment of the invention inaccordance with a second timing of the FIG. 8 timing diagram;

FIG. 11 illustrates a third shared operational amplifier configurationof two stages of the pipelined analog-to-digital converter constructedin accordance with an exemplary embodiment of the invention inaccordance with a third timing of the FIG. 8 timing diagram;

FIG. 12 illustrates comparative simulations of the conventionalpipelined analog-to-digital converter and the pipelinedanalog-to-digital converter constructed in accordance with an exemplaryembodiment of the invention;

FIGS. 13 a-c illustrate a portion of a two-channel signal processingcircuit that shares operational amplifiers between respective portionsof the channels constructed in accordance with an exemplary embodimentof the invention;

FIG. 14 is a block diagram of a CMOS imager, which utilizes either thepipelined analog-to-digital converter or the shared channel processingcircuitry constructed in accordance with an exemplary embodiment of theinvention; and

FIG. 15 is a block diagram of a processing system utilizing the imagingsystem illustrated in FIG. 14.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 8 is an exemplary timing diagram for two stages of pipelinedanalog-to-digital converter constructed in accordance with an exemplaryembodiment of the invention. The construction of the pipelinedanalog-to-digital converter of the invention is similar to theanalog-to-digital converter 10 illustrated in FIG. 1, but with thefollowing notable exceptions. The pipelined analog-to-digital converterof the invention uses a shared operational amplifier circuit 220 (asshown in FIGS. 9-11). The circuit 220 differs from other sharedoperational amplifier circuits 120 (FIG. 5-6) because it includes adischarge switch S7 connected between an input of the operationalamplifier 32-and a ground potential. As is described below, when thedischarge switch S7 is closed (at the timing illustrated in FIG. 8), adischarge path from the operational amplifier 32 input to ground iscreated. This discharge path will discharge any parasitic capacitanceseen at the operational amplifier 32 input, which substantiallymitigates the memory effect and the problems associated with the memoryeffect.

As shown in FIG. 8, when the second clock signal PHI2 is asserted, thefirst stage STAGE 1 undergoes a sampling operation while the secondstage undergoes an amplifying operation. When the first clock signalPHI1 is asserted, the first stage STAGE 1 undergoes an amplifyingoperation while the second stage undergoes a sampling operation. Duringthe time after the first clock signal PHI1 transitions low, but beforethe second clock signal PHI2 transitions high, a short reset pulse RESETis asserted. The short reset pulse RESET is also asserted during thetime after the second clock signal PHI2 transitions low, but before thefirst clock signal PHI1 transitions high. The assertion of the clocksignals PHI1, PHI2 and the reset pulse RESET configures the circuit 220as follows.

Referring to FIG. 9, a clock generator 240 generates the first clocksignal PHI1. When the first clock signal PHI1 is asserted, switches S2,S3 and S4 are closed, while switches S1, S5, S6 and S7 are opened. Thisswitch configuration allows the stage 1 arithmetic unit 20 ₁ to amplify,using the operational amplifier 32, a signal stored in the stage 1sampling capacitor Cs. Thus, the output Vout of the amplifier 32 andthus, circuit 220, is an amplified stage 1 signal. In addition, theswitch configuration allows the stage 2 arithmetic unit 20 ₂ to store aninput analog signal Vin in the stage 2 sampling capacitor Cs.

FIG. 10 illustrates a reset/discharge operation of the circuit 220. Asis shown in FIG. 8, a reset pulse RESET is generated during two timeperiods: (1) during the time after the first clock signal PHI1transitions low, but before the second clock signal PHI2 transitionshigh; and (2) during the time after the second clock signal PH12transitions low, but before the first clock signal PHI1 transitionshigh. Referring to FIG. 10, the clock generator 240 generates thereset/discharge pulse at the appropriate timing. Since both the firstand second clock signals PHI1, PHI2 are low (i.e., not asserted),switches S1-S6 are opened. Due to the reset pulse RESET, switch S7 isclosed. The closure of switch S7 forms a discharge path from the firstinput of the operational amplifier 32 to a ground potential. Thisdischarge path allows any parasitic capacitance Cp present at the firstinput of the amplifier 32 to be discharged. Any subsequentsampling/amplification operations will not be adversely impacted byparasitic capacitance Cp, which means that the circuit 220 of theinvention, and the pipelined analog-to-digital converter utilizing thecircuit 220, will not suffer from the memory effect.

Referring to FIG. 11, the clock generator 240 generates the second clocksignal PHI2. When the second clock signal PHI2 is asserted, switches S1,S5 and S6 are closed, while switches S2, S3, S4 and S7 are opened. Thisswitch configuration allows the stage 1 arithmetic unit 20 ₁ to store aninput analog signal Vin in the stage 1 sampling capacitor Cs. Inaddition, the switch configuration allows the stage 2 arithmetic unit 20₂ to amplify, using the operational amplifier 32, a signal stored in thestage 2 sampling capacitor Cs. Thus, the output Vout of the amplifier 32and thus, circuit 220, is an amplified stage 2 signal.

It should be appreciated that the clock signals PHI1, PHI2 and the resetpulse RESET (FIG. 8) may be generated by the dock generator 240 or otherdevice, which may be configured to, or controlled to, generate the clocksignals PHI1, PHI2 and the reset pulse RESET at the timing illustratedin FIG. 8. The clock generation circuitry may be the same circuitry usedin the conventional pipelined analog-to-digital converters, with themodification of generating the reset pulse RESET at the appropriatetiming.

FIG. 12 illustrates comparative simulations of the conventionalpipelined analog-to-digital converter and the pipelinedanalog-to-digital converter constructed in accordance with an exemplaryembodiment of the invention. In FIG. 12, line 300 represents the idealamplifier 32 output. Curve 302 represents the simulated amplifieroutput, when the reset/discharging technique of the invention is used.That is, curve 302 represents the outputs achieved when the short resetpulse is applied in the time periods before the clock signals PHI1, PHI2switch the shared configuration of the circuit 220. Curve 302 indicatesthat the output is substantially at the level of the ideal output 300.By contrast, curve 304 represents the simulated amplifier output whenthe reset/discharging technique of the invention is not used. That is,curve 304 represents the prior art switching used in the sharedconfiguration of circuit 120. Curve 304 indicates that the output issubstantially above the level the ideal output 300. This is due to thememory effect described above. Accordingly, the present invention is avast improvement of today's conventional pipelined analog-to-digitalconverters that share amplifiers among pipeline stages.

One of the advantages of using the short reset pulse RESET (FIG. 8) todischarge any parasitic capacitance at the operational amplifier 32input includes reducing the residual error associated with the parasiticcapacitance, especially when the analog-to-digital converter isover-driven. Another advantage includes reducing the worst case slewingtime of the circuit 220. With the invention, the maximum output swing ofthe operational amplifier 32 is a reference voltage Vref. Without thereset/discharge feature of the invention, the worst case output swing ofthe operational amplifier could be at least twice the reference voltageVref.

FIGS. 13 a-c illustrate a portion of a two-channel signal processingcircuit 250 that shares operational amplifiers 32 between respectiveportions of the channels 252 _(a), 252 _(b) constructed in accordancewith an exemplary embodiment of the invention. That is, the circuit 250utilizes the short reset pulse to discharge any capacitance seen at theshared operational amplifier 32. The first channel 252 _(a) comprisesfour capacitors C1, C2, C3, C4 and switches S1, S2, S3, S4, S5, S6, S13,S14, S15, S16, S17, S18. The second channel 252 b comprises fourcapacitors C5, C6, C7, C8 and switches S7, S8, S9, S10, S11, S12, S19,S20, S21, S22, S23, S24. Between the two channels 252 _(a), 252 _(b), isan operational amplifier 32 and two discharge switches S25, S26. Thecircuitry operates at the timing illustrated in FIG. 8.

FIG. 13 a illustrates the configuration for the two channels 252 _(a),252 _(b) when the first clock signal PHI1 is asserted. When the firstclock signal PHI1 is asserted, switches S1, S2, S5, S6, S13 and S14 areclosed in the first channel 252 _(a) while switches S11, S12, S19 andS20 are closed in the second channel 252 _(b). This connects the firstchannel 252 _(a) to receive differential input signals Vinp, Vinn. Thesecond channel 252 _(b) amplifies signals previously stored incapacitors C5-C8. Vcm is a common mode voltage used to place charge on(or read charge out o f the capacitors C1-C8.

FIG. 13 b illustrates the configuration for the two channels 252 _(a),252 _(b) when the reset pulse RESET is asserted. That is, FIG. 13 billustrates the discharge operation. When the reset pulse RESET isasserted, only switches S25 and S26 are closed. Since both the first andsecond clock signals PHI1, PHI2 are low (i.e., not asserted), switchesS1-S24 are opened. The closure of switch S25 forms a discharge pathbetween the inputs of the operational amplifier 32. This discharge pathallows any parasitic capacitance present at the amplifier 32 to bedischarged. Any subsequent sampling/amplification operations will not beadversely impacted by parasitic capacitance, which means that the twochannels 252 _(a), 252 _(b) will not suffer from the memory effect. FIG.13 b illustrates the closure of output switch S26 during the dischargeoperation, which may help mitigate the memory effect at the amplifier 32outputs. It should be appreciated, however, that switch S26 (and theclosure of switch 26) is not required to practice the invention.

FIG. 13 c illustrates the configuration for the two channels 252 _(a),252 _(b) when the second clock signal PHI2 is asserted. When the secondclock signal PHI2 is asserted, switches S15, S16, S17 and S18 are closedin the first channel 252 _(a) while switches S7, S8, S9, S10, S22 andS23 are closed in the second channel 252 _(b). This connects the secondchannel 252 _(b) to receive differential input signals Vinp, Vinn. Thefirst channel 252 _(a) amplifies signals previously stored in capacitorsC1-C4.

FIG. 14 illustrates an exemplary imager 700 that may utilize theanalog-to-digital converter or shared channel processing circuitryconstructed in accordance with the invention. The Imager 700 has a pixelarray 705 comprising rows and columns of pixels. Row lines areselectively activated by a row driver 710 in response to row addressdecoder 720. A column driver 760 and column address decoder 770 are alsoincluded in the imager 700. The imager 700 is operated by the timing andcontrol circuit 750, which controls the address decoders 720, 770. Thecontrol circuit 750 also controls the row and column driver circuitry710, 760.

A sample and hold circuit 761 associated with the column driver 760reads a pixel reset signal Vrst and a pixel image signal Vsig forselected pixels. A differential signal (Vrst−Vsig) is amplified bydifferential amplifier 762 for each pixel and is digitized by thepipelined analog-to-digital converter 775 of the invention. Theanalog-to-digital converter 775 supplies the digitized pixel signals toan image processor 780, which forms a digital image. Alternatively, thesample and hold circuit 761 and the analog-to-digital converter 775 maybe connected in a shared two channel configuration such as theconfiguration illustrated in FIGS. 7 c and 13 a-13 c. Each channel wouldbe responsible for a different set of pixel signals (e.g., one channelcan process red and blue pixel signals, while the other channelprocesses green pixel signals).

FIG. 15 shows a system 1000, a typical processor system modified toinclude an imaging device 1008 (such as the imaging device 700illustrated in FIG. 14) of the invention. The processor system 1000 isexemplary of a system having digital circuits that could include imagesensor devices. Without being limiting, such a system could include acomputer system, camera system, scanner, machine vision, vehiclenavigation, video phone, surveillance system, auto focus system, startracker system, motion detection system, image stabilization system, anddata compression system.

System 1000, for example a camera system, generally comprises a centralprocessing unit (CPU) 1002, such as a microprocessor, that communicateswith an input/output (I/O) device 1006 over a bus 1020. Imaging device1008 also communicates with the CPU 1002 over the bus 1020. Theprocessor-based system 1000 also includes random access memory(RAM)1004, and can include removable memory 1014, such as flash memory,which also communicate with the CPU 1002 over the bus 1020. The imagingdevice 1008 may be combined with a processor, such as a CPU, digitalsignal processor, or microprocessor, with or without memory storage on asingle integrated circuit or on a different chip than the processor.

The processes and devices described above illustrate preferred methodsand typical devices of many that could be used and produced. The abovedescription and drawings illustrate embodiments, which achieve theobjects, features, and advantages of the present invention. However, itis not intended that the present invention be strictly limited to theabove-described and illustrated embodiments. Any modification, thoughpresently unforeseeable, of the present invention that comes within thespirit and scope of the following claims should be considered part ofthe present invention.

1. A method of operating a pipelined analog-to-digital convertercomprising first and second stages sharing an amplifier, said methodcomprises: performing a first discharge operation at the amplifier inresponse to a first reset pulse during a first period in which the firstand second stages change operations, wherein the first period occursafter a first clock signal is used to cause the first stage to perform afirst operation and the second stage to perform a second operation, butbefore a second clock signal is used to cause the first stage to performthe second operation and the second stage to perform the firstoperation; and performing a second discharge operation at the amplifierduring a second period in which the first and second stages changeoperations, wherein the second period occurs after the second clocksignal is used to cause the first stage to perform the second operationand the second stage to perform the first operation.
 2. An imagercomprising: a pixel array; sample and hold circuitry coupled to receiveanalog signals from pixels within the array; an amplification circuitfor amplifying the analog signals; and a pipelined analog-to-digitalconverter connected to receive and convert the amplified analog signalsto digital signals, said converter comprising: first and second pipelinestages and a discharge circuit, the first and second pipeline stages areadapted to be switchably connected to an amplifier in response to firstand second clock signals, the discharge circuit causing a dischargeoperation to occur at an input of the amplifier in response to a resetpulse during a first period after the first clock signal, but before thesecond clock signal, in which the connections between the first andsecond pipeline stages and the amplifier switch, the discharge circuitcausing another discharge operation to occur at the input of theamplifier in response to another reset pulse during a second periodafter the second clock signal, but before the first clock signal, inwhich the connections between the first and second pipeline stages andthe amplifier switch.
 3. The imager of claim 2, wherein said dischargeoperation forms a parasitic capacitance discharge path from the input ofthe amplifier to a discharge area.
 4. The imager of claim 2, whereinsaid discharge circuit comprises a switch coupled between a groundpotential and the input the amplifier, wherein the switch is adapted toreceive the reset pulse between the first and second clock signals andconnect the input of the amplifier to the ground potential in responseto the reset pulse.
 5. The imager of claim 2, wherein theanalog-to-digital converter further comprises a clock generator, saidclock generator generating non-overlapping first and second clocksignals and a plurality of reset pulses, said clock generator applyingthe first and second clock signals and the reset pulses to the pipelinestages and the discharge circuit to change the connections between thepipeline stages, discharge circuit and the amplifier.
 6. The imager ofclaim 5, wherein said first clock signal causes said first stage to havea first connection to the amplifier and to perform a first operation andsaid second stage to have a second connection to the amplifier and toperform a second operation.
 7. The imager of claim 6, wherein saidsecond clock signal causes said first stage to have the secondconnection to the amplifier and to perform the second operation and saidsecond stage to have the first connection to the amplifier and toperform the first operation.
 8. The imager of claim 5, wherein saidreset pulses cause the discharge circuit to have a connection between adischarge path and the input of the amplifier and to perform thedischarge operation.
 9. The imager of claim 5, wherein the reset pulsesare generated after the first clock signal transitions from a firststate to a second state and after the second clock signal transitionsfrom the first state to the second state.
 10. An imager comprising: apixel array; a first signal processing channel coupled to receive firstanalog signals from pixels within the array; a second signal processingchannel coupled to receive second analog signals from pixels within thearray; amplifier adapted to be switchably connected between the firstand second channels; and a discharge circuit adapted to be switchablyconnected to an input of the amplifier, the discharge circuit causing adischarge operation to occur at an input of the amplifier during aperiod in which the connections between the first and second channelsand the amplifier switch, the discharge circuit comprising a switchcoupled between a ground potential and the input of the amplifier,wherein the switch is adapted to receive a reset pulse and connect theinput of the amplifier to the ground potential in response to the resetpulse.
 11. The imager of claim 10, wherein the discharge operation formsa parasitic capacitance discharge path from the input of the amplifierto a discharge area.
 12. The imager of claim 10, wherein theanalog-to-digital converter further comprises a clock generator, saidclock generator generating non-overlapping first and second clocksignals and a plurality of reset pulses, said clock generator applyingthe first and second clock signals and the reset pulses to the channelsand the discharge circuit to change the connections between thechannels, discharge circuit and the amplifier.
 13. The imager of claim12, wherein said first clock signal causes said first channel to have afirst connection to the amplifier and to perform a first operation andsaid second channel to have a second connection to the amplifier and toperform a second operation.
 14. The imager of claim 13, wherein saidsecond clock signal causes said first channel to have the secondconnection to the amplifier and to perform the second operation and saidsecond channel to have the first connection to the amplifier and toperform the first operation.
 15. The imager of claim 12, wherein saidreset pulses cause the discharge circuit to have a connection between adischarge path and the input of the amplifier and to perform thedischarge operation.
 16. The imager of claim 12, wherein the resetpulses are generated after the first clock signal transitions from afirst state to a second state and after the second clock signaltransitions from the first state to the second state.